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Copyright © LAMSIM Enterprises Inc.
White Paper
Split Planes and What Happens When
Microstrip Signals Cross Them
Issue 01
Bert Simonovich
12/9/2017
THIS DOCUMENT IS PROVIDED FOR INFORMATIONAL PURPOSES ONLY. THE
AUTHOR(S) MAKE NO REPRESENTATIONS ABOUT THE SUITABILITY OF THE
INFORMATION CONTAINED HEREIN FOR ANY PURPOSE. THIS DOCUMENT IS
PROVIDED "AS IS" WITHOUT WARRANTY OF ANY KIND.
©LAMSIM Enterprises Inc.
3
Split Planes and What Happens When
Microstrip Signals Cross Them
When discussing signal integrity (SI) issues there is always a great debate when signals on one
layer of a printed circuit board (PCB) crossing over split or a slot in the reference planes on an
adjacent layer. On the one hand, some argue that crossing a split plane should never be done
because of the increased risk in crosstalk and possible failure to pass electromagnetic
compatibility (EMC) compliance. On the other hand, others stressed that if the width of the gap
and power/ground layers in the stackup were engineered carefully, this may not be as big of an
issue. So who’s right?
Well, like all things involving signal integrity, the answer is, “it depends”. And the best way to
answer “it depends” is to put in the numbers. This white paper attempts to dispel some of the
myths about signals crossing split planes.
To start off with let us look at a typical 4 layer PCB ~ 62 mils thick with a stackup shown in
Figure 1. The outer two layers are microstrip signal layers and the inner two layers are power and
ground. The trace widths are 7 mils wide with 8 mil separation. When driven differentially the
impedance is ~100 ohms; and when driven signal-ended (SE), the impedance is ~ 56 ohms.
Figure 1 Simple 4 layer PCB stackup
It is common nowadays to have multiple power rails in modern designs. On a 4-layer board this
means the power layer, more often than not, will be split up and as a result, traces crossing splits
or slots on adjacent reference planes are often unavoidable.
Let’s assume we have a pair of traces on the top layer crossing a 50 mil gap on the adjacent layer
as shown in Figure 2. The cross-section of the microstrip sections before and after the gap sees
the dielectric thickness (H1) from the top layer to the power reference plane. Because the gap
©LAMSIM Enterprises Inc.
4
section has no reference plane on the adjacent power layer, the next reference plane is the GND
layer adjacent to the bottom layer. As a result, the dielectric thickness across the gap equals the
thickness of H1 plus the thickness of the 1 oz. power layer (t2) plus the thickness of the next
dielectric layer (H2). If the thickness of the 1 oz. power layer is 1.2 mils, then the total thickness
of dielectric is 51.2 mils across the gap.
A first order approximation of this topology is a combination of three transmission line segments
with two different impedances. The first and last segments are 100 ohms differential and 56
ohms SE, while the trace impedances across the gap are ~134 ohms differential and ~103 ohms
SE. Since the impedance across the gap is higher than the first and last segments, we expect to
see be a positive reflection over the length of the gap. The height and width of the reflection will
be a function of the rise-time and geometry of the gap. A fast rise-time with a long gap will give
a higher reflection than a slow rise-time and a short gap.
Figure 2 Cross-sectional geometries relative to gap model topology
To see just how much of an issue this is we can quickly model and simulate it with Keysight
ADS [1] as shown in Figure 3. Two transmission line segments before and after the gap section
were modeled with internal 2D field solver using the “TLines-Line Type” pallet. The gap section
was modeled and simulated with Momentum 3D planar field solver in order to properly capture
the electromagnetic effects as the signals cross the gaps. Both shared the same substrate
definition. The S-parameter results from Momentum were saved in touchstone format and
brought back into the ADS schematic.
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